The present invention relates to a high frequency filter, in particular, relates to a novel structure of a bandpass filter of dielectric waveguide type, which is suitable for use especially in the range from the VHF bands to the comparatively low frequency microwave bands. The present filter relates particularly to such a filter having a plurality of resonator rods each coupled electrically and/or magnetically with the adjacent resonators, and can be conveniently installed in a mobile communication system.
Such kind of filters must satisfy the requirements that the size is small, the energy loss in a high frequency is small, the manufacturing process is simple, and the characteristics are stable.
When a filter is composed of a plurality of elongated rod resonators, the size of each resonator and the coupling between resonators must be considered.
First, three prior filters for the use of said frequency bands will be described.
FIG. 1A shows the perspective view of a conventional interdigital filter, which has been widely utilized in the VHF bands and the low frequency microwave bands. In the figure, the reference numerals 1-1 through 1-5 are resonating rods which are made of conductive material, 2-1 through 2-4 are gaps between adjacent resonating rods, and 3 is a case. The 3-1 through 3-3 are conductive walls of said case 3. A cover 3-4 of the case 3 is not shown for the sake of the simplicity of the drawing. A pair of exciting antennas 4 are provided for the coupling of the filter with an external circuit. The length of each illustrated resonating rod 1-1 through 1-5 is selected as to be substantially equivalent to one quarter of a wavelength, and one end of the resonating rods are short-circuited alternately to the confronting conductive walls 3-1 and 3-2, while the opposite ends thereof are free standing.
As is well known, when a resonator stands on a conductive plane, a magnetic flux distributes so that the density of the magnetic flux is maximum at the foot of the resonator, and is zero at the top of the resonator, while the electrical field distributes so that said field is maximum at the top of the resonator and the field at the foot of the resonator is zero. Therefore, when a pair of resonators are mounted on a single conductive plane, those resonators are coupled with each other magnetically and electrically, and the magnetic coupling is performed at the foot of the resonators, and the electrical coupling is performed at the top of the resonators. However, since the absolute value of the magnetic coupling is the same as that of the electrical coupling, and the sign of the former is opposite to the latter, the magnetic coupling is completely cancelled by the electrical coupling, and as a result, no coupling is obtained between two resonators.
In order to solve that problem, an interdigital filter arranges the resonators alternately on a pair of confronting conductive walls. In that case, the two adjacent resonators are electrically coupled with each other as shown in FIG. 1B, where the magnetic flux M which has the maximum value at the foot of the resonator does not contribute to the coupling of the two resonators since the foot of the first resonator 1-1 located far from the foot of the second resonator 1-2, and so, only the electrical field E contributes to the coupling of the two resonators.
However, said interdigital filter has the disadvantage that the manufacture of the filter is cumbersome and subsequently the filter is costly, since each of the resonating rods are fixed alternately to the confronting two conductive walls to obtain a high enough coupling coefficient between each of the resonating rods.
FIG. 2 shows the perspective view of another conventional filter, which is called a comb-line type filter, and has been utilized in the VHF bands and the low frequency microwave bands. In the figure, the reference numerals 11-1 through 11-5 are conductive resonating rods with one end thereof left free standing while opposite end thereof short-circuited to the single conductive wall 13-1 of a conductive case 13. The length of each resonating rod 11-1 through 11-5 is selected to be a little shorter than a quarter of a wavelength. The resonating rod acts as inductance (L), and capacitance (C) is provided at the head of each resonating rod for providing the resonating condition. In FIG. 2, said capacitance is accomplished by the dielectric disk 11a-1 through 11a-5 and the conductive bottom wall 13-2 of the case 13. The gaps 12-1 through 12-4 between each of the resonating rods, and the capacitance between the dielectric disks 11a-1 through 11a-5, and the bottom wall 13-2 provide the necessary coupling between each of the resonating rods. A pair of antennas 14 are provided for the coupling between the filter and external circuits.
With this type of filter, the resonating rods 11-1 through 11-5 are fixed on the single bottom wall 13-1 and the manufacturing cost can be reduced as far as this point is concerned, but there is the shortcoming in that the manufacture of the capacitance (C) with an accuracy of, for instance, several %, is rather difficult, resulting in no cost merit. Therefore, the advantage of a comb-line type filter is merely that it can be made smaller than an interdigital filter.
Further, although we try to shorten the resonators in the filters of FIG. 1A and/or FIG. 2 by filling dielectric material in a housing, it is almost impossible since the structure of the filters are complicated. It should be noted that the material of the dielectric body for the use of a high frequency filter is ceramics for obtaining the small high frequency loss, and it is difficult to manufacture the ceramics with the complicated structure to cover the interdigital electrodes of FIG. 1A, or the combination of the disks and the rods of FIG. 2. If we try to fill the housing with plastics, the high frequency loss by plastics would be larger than the allowable upper limit.
Further, a dielectric filter which has a plurality of dielectric resonators has been known. However, a dielectric filter has the shortcoming that the size of each resonator is rather large even when the dielectric constant of the material of the resonators is the largest possible.
Accordingly, the present applicant has proposed the filter having the structure of FIG. 3A (U.S. Ser. No. 92,670, now U.S. Pat. Nos. 4,283,697, and 37,419, now U.S. Pat. No. 4,255,729, Canadian application 339,477, GB serial number 7940057, West Germany P2946 836.8, France 79 28588, Holland 7908381, Sweden 7909547-7, Canada, 326,986, and EPC 79101456.6). In FIG. 3A, each resonator has a circular center conductor (31-1 through 31-5), and the cylindrical dielectric body (31a-1 through 31a-5) covering the related center conductor, and each of the resonators are fixed on the single conductive plane 33-1 of the housing 33, leaving the air gaps (32-1 through 32-4) between the resonators. The 34 are antennas for coupling the filter with external circuits. The case 33 has the closed conductive walls having the walls 33-1, 33-2 and 33-3 (upper cover wall is not shown). The structure of the filter of FIG. 3A has the advantage that the length L of a resonator is shortened due to the presence of the dielectric body covering the conductor, and the resonators are coupled with each other although the resonators are fixed on a single conductive plane due to the presense of the dielectric bodies covering the center conductors.
When the two resonators contact with each other as shown in FIG. 3B, those resonators do not couple with each other, because the electrical coupling between the two resonators is completely cancelled by the magnetical coupling between the two resonators. In this case, the dielectric covering 31-1 and 31-2 do not contribute to the coupling between the resonators. On the other hand, when an air space 32-1 is provided between the surfaces of the dielectric bodies 31-1 and 31-2 as shown in FIG. 3C, some electric field (p) originated from one resonator is curved at the surface of the dielectric body (the border between the dielectric body and the air), due to the difference of the dielectric constants of the dielectric body 31-1 or 31-2, and the air, so that the electric field is directed to an upper or bottom conductive wall. That is to say, the electric field (p) leaks, and the electrical coupling between the two resonators is decreased, and so that decreased electrical coupling can not cancell all the magnetic coupling which is not affected by the presence of the dielectric cover. Accordingly, the two resonators are coupled magnetically by the amount equal to the decrease of the electrical coupling. That decrease of the electrical coupling is caused by the leak of the electrical field at the border between the dielectric surface and the air, due to the presence of the air gap 32-1.
The leak of the electric field to an upper and/or bottom conductive wall increases with the length (x) between the two resonators, or the decrease of the electrical coupling increases with that length (x). Therefore, the overall coupling between resonators which is the difference between the magnetic coupling and the electrical coupling increases with the length (x) so long as that value (x) is smaller than the predetermined value (x.sub.0). When the length (x) exceeds that value (x.sub.0), the absolute value of both the electrical coupling and the magnetic coupling becomes small, and so the total coupling decreases with the length (x).
However, we found that the filter of FIG. 3A has the disadvantage that the leak (p) of the electrical field to an upper and/or bottom wall is considerably affected by the manufacturing error of both the housing and the dielectric cover. That is to say, the small error of the gap between the upper and/or bottom wall and the dielectric cover, and/or the small error of the size of the dielectric cover provides much error for the characteristics of the filter. Further, the filter is sometimes unstable since the resonators are fixed only at one end of them.
Further, we found that the coupling coefficient between resonators is not enough for providing a wideband filter.